Communication Concepts AN779 Application Note - Manual (Page 5)

AR
C
HIVE INF
O
RMA
TI
O
N
PRODUCT TRANSFERRED T
O
M/A
–
COM
AN779
5 RF Application Reports
2 3 5 7 10 15 20
f, FREQUENCY (MHz)
INPUT
VSWR
2.0
1.5
1.0
40
35
30
1.5 30
(%)
η
VSWR
η
Figure 7. Input VSWR and Combined Collector
Efficiency of Both Stages
20 W, 55 dB HIGH PERFORMANCE DRIVER
12–Volt Version
The second amplifier employs the MHW591 hybrid
module to drive a pair of larger devices which can be
operated Class A or AB, depending on the requirements.
Transistors such as MRF449 and MRF455 are recom-
mended for Class A and MRF433 for Class AB operation.
A 24–28 volt version with MHW592 and a pair of MRF401s
was also designed, and some of the test data will be
presented. For Class A, the power parts should be replaced
with MRF426s.*
These amplifiers are a good example of how a good gain
flatness can be achieved across the four–octave band, with
simple RC input networks and negative feedback, while
maintaining a reasonable input VSWR.
The MHW591 is employed as a predriver in this unit. The
MHW591 and its counterpart, MHW592, were developed for
low–level SSB driver applications from 1.0 MHz to 250 MHz.
The Class A operation results in a steady–state current drain
of approximately 0.32 A, which does not vary with the signal
level. At an output level of 600 mW PEP, the IMD is typically
better than –40 dB, which can be considered sufficiently
good for most purposes. Since the power gain is specified
as 36.5 dB, the maximum drive level for the 600 mW output
is 0.13 mW, or –9 dBM. For the final power output of 20 W,
a power gain of 15.2 dB minimum is required at the highest
operating frequency for the power transistors. A good,
inexpensive device for this is the MRF433, which has a 20 dB
minimum gain and –30 dB IMD specification at an output
level of 12.5 W PEP. The push–pull configuration, due to
inconsistent ground planes and broadbanding due to
matching compromises usually results in 2 dB to 3 dB gain
losses from figures measured in a test fixture. Assuming a
transistor power gain of 18 dB, the total will be 54.5 dB,
representing an input power of –11 dBM. Later measure-
ments, however, indicated a gain of 56 dB (
± 0.5 dB) at the
specified power output, making the input level around
–13 dBM.
Biasing and Feedback
The bias circuit employed with this amplifier is basically
similar to the one described earlier, with the exception of
having an emitter follower output. A second diode in series
with the one normally seen with the clamping diode method
compensates for the voltage drop in the base–emitter
junction of the emitter follower, Q1 (Figure 8). The minimum
current through D1 and D2 is (I
C
/h
FE
) (Q2 + Q3)/h
FE
(Q1),
and in this case (2.5/40)/40 = 1.5 mA. Typical h
FE
for the
MRF433 is 40, and with the devices biased to 200 mA each,
the standby base current is 10 mA. In operation the load
current of Q1 then varies between 10 and 62 mA. A Case
77 transistor exhibiting low variations in base–emitter
saturation voltage over this current range is MJE240.
Base–emitter saturation voltage determines the bias source
impedance, which should not exceed approximately
0.3 ohm, representing a 20 mV variation in voltage from idle
to full drive conditions. If source impedance exceeds
0.3 ohms, a capacitor of 500–1000
µF should be connected
from the emitter of Q1 to ground.
The peak dissipation of Q2 is under one watt, making it
possible to mount the transistor directly to the circuit board
without requiring any additional heat sinking.
Diodes D1 and D2 are located on the lower side of the
board, near Q2 and Q3 (Figure 9). The leads are formed
to allow the diodes to come into contact with the transistor
flanges. The thermal contact achieved in this manner is not
the best possible, even when the gaps are filled with silicone
compound, but the thermal time constant is lower than with
most other methods. Both diodes are used for temperature
tracking, although the voltage drop of only one is required
to compensate for the V
BE
forward drop of Q1. The
advantages of this circuit are simplicity, low standby current
drain, and ease of adjustment with a small trimpot.
The voltages for the negative feedback are derived
separately from the collectors of Q2 and Q3 through L6, R6
and L7, R7. Capacitors C5 and C6 are used for dc isolation.
Because of the high RF voltage levels on the collectors, this
method is only feasible in low– and medium–power
amplifiers. At higher power levels, the power–handling
requirements for the series resistors (Figure 8), which must
be noninductive, become impractical. A feedback voltage
source with lower impedance must be provided in such
cases.
8
The MRF433 has a higher figure of merit (emitter
periphery/base area) than the MRF475, for example. This
results in smaller differences in power gain per given
bandwidth, since the device is operating farther away from
the 6 dB/octave slope.
9 Disregarding the package induc-
tances, which affect the Q, the higher figure of merit makes
such devices more suitable for broadband operation. The
2 MHz and 30 MHz
∆G
PE
of the MRF433 is 8 dB, which
is divided equally between the negative feedback and the
leveling networks C3, R4 and C4, R5. The 2 MHz and 30
MHz impedance values are 9.1, –j3.5 and 2.5, –j2.2 ohms,
respectively, although the 2 MHz values are not given in the
data sheet.
At 30 MHz we can first determine what type of transformer
is needed for the 50–ohm input interface. The effective
transformer load impedance is 2
√(2.5 2 + 2.2 2 ) + 2 (2.4) ohms
(leveling networks) = 11.5 ohms, which indicates that 4:1
impedance ratio is the closest possible (see Figures 3B, 4A,
and 8). These values are accurate for practical purposes,
but they are not exact, since part of the capacitive reactance
in C3 and C4 will be cancelled, depending on the transformer
characteristics.
*To be introduced.